Methods for sensing memory elements in semiconductor devices

ABSTRACT

A memory device that, in certain embodiments, includes a plurality of memory elements connected to a bit-line and a delta-sigma modulator with a digital output and an analog input, which may be connected to the bit-line. In some embodiments, the delta-sigma modulator includes a circuit with first and second inputs and an output. The circuit is configured to combine (add or subtract) input signals. The first input may be connected to the analog input. The delta-sigma modulator may also include an integrator connected to the output of the circuit, an analog-to-digital converter with an input connected to an output of the integrator and an output connected to the digital output, and a digital-to-analog converter with an input connected to the output of the analog-to-digital converter and an output connected to the second input of the circuit.

CROSS-REFERENCE TO RELATED APPLICATION

This application is a divisional of U.S. patent application Ser. No.13/486,535, which was filed on Jun. 1, 2012, now U.S. Pat. No.8,582,375, which issued on Nov. 12, 2013, which was a divisional of U.S.patent application Ser. No. 12/951,997, which was filed on Nov. 22,2010, now U.S. Pat. No. 8,194,477, which issued on Jun. 5, 2012, whichis a divisional of U.S. patent application Ser. No. 11/820,003, whichwas filed on Jun. 15, 2007, now U.S. Pat. No. 7,839,703, which issued onNov. 23, 2010.

BACKGROUND

1. Field Of The Invention

Embodiments of the present invention relate generally to electronicdevices and, more specifically, to subtraction circuits anddigital-to-analog converters for delta-sigma modulators in electronicdevices.

2. Description Of The Related Art

Generally, memory devices include an array of memory elements andassociated sense amplifiers. The memory elements store data, and thesense amplifiers read the data from the memory elements. To read data,for example, a current is passed through the memory element, and thecurrent or a resulting voltage is measured by the sense amplifier.Conventionally, the sense amplifier measures the current or voltage bycomparing it to a reference current or voltage. Depending on whether thecurrent or voltage is greater than the reference, the sense amplifieroutputs a value of one or zero. That is, the sense amplifier quantizesor digitizes the analog signal from the memory element into one of twologic states.

Many types of memory elements are capable of assuming more than twostates. That is, some memory elements are capable of multi-bit storage.For instance, rather than outputting either a high or low voltage, thememory element may output four or eight different voltage levels, eachlevel corresponding to a different data value. However, conventionalsense amplifiers often fail to distinguish accurately between theadditional levels because the difference between the levels (e.g., avoltage difference) in a multi-bit memory element is often smaller thanthe difference between the levels in a single-bit memory element. Thus,conventional sense amplifiers often cannot read multi-bit memoryelements reliably. This problem may be increased as high performancemulti-bit memory elements become increasingly dense, thereby reducingthe size of the memory elements and the difference between the levels(e.g., voltage) to be sensed by the sense amplifiers.

A variety of factors may tend to prevent the sense amplifier fromdiscerning small differences in the levels of a multi-bit memoryelement. For instance, noise in the power supply, ground, and referencevoltage may cause an inaccurate reading of the memory element. The noisemay have a variety of sources, such as temperature variations, parasiticsignals, data dependent effects, and manufacturing process variations.This susceptibility to noise often leads a designer to reduce the numberof readable states of the memory element, which tends to reduce memorydensity and increase the cost of memory.

Conventional sense amplifiers present similar problems in imagingdevices. In these devices, an array of light sensors output a current orvoltage in response to light impinging upon the sensor. The magnitude ofthe current or voltage typically depends upon the intensity of thelight. Thus, the capacity of the sense amplifier to accurately convertthe current or voltage into a digital signal may determine, in part, thefidelity of the captured image. Consequently, noise affecting the senseamplifier may diminish the performance of imaging devices.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 illustrates an electronic device in accordance with an embodimentof the present invention;

FIG. 2 illustrates a memory device in accordance with an embodiment ofthe present invention;

FIG. 3 illustrates a memory array in accordance with an embodiment ofthe present invention;

FIG. 4 illustrates a memory element in accordance with an embodiment ofthe present invention;

FIG. 5 illustrates I-V traces of memory elements storing differentvalues, in accordance with an embodiment of the present invention;

FIG. 6 illustrates noise in the bit-line current during a readoperation;

FIG. 7 illustrates a quantizing circuit in accordance with an embodimentof the present invention;

FIG. 8 illustrates a delta-sigma sensing circuit in accordance with anembodiment of the present invention;

FIGS. 9 and 10 illustrate current flow during operation of thequantizing circuit of FIG. 8;

FIGS. 11-13 illustrate voltages in the quantizing circuit of FIG. 8 whensensing small, medium, and large currents, respectively;

FIG. 14 is a graph of bit-line current versus counter output for thequantizing circuit of FIG. 8;

FIG. 15 is a graph of count versus quantizing circuit output inaccordance with an embodiment of the present invention;

FIG. 16 is a block diagram of a delta-sigma modulator in accordance withan embodiment of the present invention;

FIG. 17 is a block diagram of a one-bit delta-sigma modulator inaccordance with an embodiment of the present invention;

FIG. 18 illustrates an example of an adder in accordance with anembodiment of the present invention;

FIGS. 19 and 20 each illustrate respective examples of avoltage-to-current converter in accordance with an embodiment of thepresent invention;

FIGS. 21-22 each illustrate respective examples of the adder illustratedby FIG. 18 in accordance with embodiments of the present invention;

FIG. 23 illustrates a second example of an adder in accordance with anembodiment of the present invention;

FIG. 24 illustrates a third example of a voltage-to-current converter inaccordance with an embodiment of the present invention;

FIGS. 25-27 each illustrate respective examples of a current switch inaccordance with an embodiment of the present invention;

FIGS. 28-31 each illustrate respective examples of a reference currentsource in accordance with an embodiment of the present invention; and

FIG. 32 is an example of a system that includes the memory device ofFIG. 2 in accordance with an embodiment of the present invention.

DETAILED DESCRIPTION OF SPECIFIC EMBODIMENTS

Various embodiments of the present invention are described below. In aneffort to provide a concise description of these embodiments, not allfeatures of an actual implementation are described in the specification.It should be appreciated that in the development of any such actualimplementation, as in any engineering or design project, numerousimplementation-specific decisions must be made to achieve thedevelopers' specific goals, such as compliance with system-related andbusiness-related constraints, which may vary from one implementation toanother. Moreover, it should be appreciated that such a developmenteffort might be complex and time consuming but would nevertheless be aroutine undertaking of design, fabrication, and manufacture for those ofordinary skill having the benefit of this disclosure.

Some of the subsequently described embodiments may address one or moreof the problems with conventional sense amplifiers discussed above. Someembodiments include a quantizing circuit configured to detect smalldifferences in voltages and/or currents. As explained below, thequantizing circuit may sample the measured electrical parameter onmultiple occasions and filter, e.g., average or sum, the samples toreduce the impact of noise. As a result, in some embodiments, thequantizing circuit may resolve small differences between voltage orcurrent levels in multi-bit memory elements and/or light sensors, whichmay allow circuit designers to increase the number of bits stored permemory element and/or the sensitivity of an imaging device.

The following description begins with an overview of examples of systemsthat employ quantizing circuits in accordance with embodiments of thepresent invention, and the problems within these systems that may beaddressed by the quantizing circuits as described with reference toFIGS. 1-7. Then, specific examples of a quantizing circuit are describedwith reference to FIGS. 8-15. Next, block diagrams of delta-sigmamodulators that may be employed in a quantizing circuit are describedwith reference to the FIGS. 16 and 17. Finally, circuits for subtractingvoltages in a delta-sigma modulator are described with reference toFIGS. 18-25, and circuits for supplying a reference current to adelta-sigma modulator are described with reference to the FIGS. 26-29.

FIG. 1 depicts an electronic device 10 that may be fabricated andconfigured in accordance with one or more of the present embodiments.The illustrated electronic device 10 includes a memory device 12 that,as explained further below, may include multi-bit memory elements andquantizing circuits. Alternatively, or additionally, the electronicdevice 10 may include an imaging device 13 having the quantizingcircuits.

Myriad devices may embody one or more of the present techniques. Forexample, the electronic device 10 may be a storage device, acommunications device, an entertainment device, an imaging system, or acomputer system, such as a personal computer, a server, a mainframe, atablet computer, a palm-top computer, or a laptop.

FIG. 2 depicts a block diagram of an embodiment of the memory device 12.The illustrated memory device 12 may include a memory array 14, aquantizing circuit 16, a column decoder 18, a column address latch 20,row drivers 22, a row decoder 24, row address latches 26, and controlcircuitry 28. As described below with reference to FIG. 3, the memoryarray 14 may include a matrix of memory elements arranged in rows andcolumns. As will be appreciated, the imaging device 13 (FIG. 1) mayinclude similar features except that in the case of an imaging device13, the memory array 14 will include a matrix of imaging elements, suchas complementary-metal-oxide semiconductor (CMOS) imaging elements.

When accessing the memory elements, the control circuitry may receive acommand to read from or write to a target memory address. The controlcircuitry 28 may then convert the target address into a row address anda column address. In the illustrated embodiment, the row address bus 30transmits the row address to the row address latches 26, and a columnaddress bus 32 transmits column address to the column address latches20. After an appropriate settling time, a row address strobe (RAS)signal 34 (or other controlling clock signal) may be asserted by thecontrol circuitry 28, and the row address latches 26 may latch thetransmitted row address. Similarly, the control circuitry 28 may asserta column address strobe 36, and the column address latches 20 may latchthe transmitted column address.

Once row and column addresses are latched, the row decoder 24 maydetermine which row of the memory array 14 corresponds to the latchedrow address, and the row drivers 22 may assert a signal on the selectedrow. Similarly, the column decoder 18 may determine which column of thememory array 14 corresponds with the latched column address, and thequantizing circuit 16 may sense a voltage or current on the selectedcolumn. Additional details of reading and writing are described below.

FIG. 3 illustrates an example of a memory array 14. The illustratedmemory array 14 includes a plurality of bit-lines 38, 40, 42, 44, and 46(also referred to as BL0-BL4) and a plurality of word-lines 48, 50, 52,54, 56, 58, 60, and 62 (also referred to as WL0-WL7). These bit-linesand word-lines are electrical conductors. The memory array 14 furtherincludes a plurality of memory elements 64, each of which may bearranged to intersect one of the bit-lines and one of the word-lines. Inother embodiments, imaging elements may be disposed at each of theseintersections. The memory elements and imaging elements may be referredto generally as internal data storage locations, i.e., devicesconfigured to convey data, either stored or generated by a sensor, whenaccessed by a sensing circuit, such as the quantizing circuits discussedbelow. The internal data storage locations may be formed on anintegrated semiconductor device that also includes the other componentsof the memory device 12 (or imaging device 13).

In some embodiments, the illustrated memory elements 64 are flash memorydevices. The operation of the flash memory elements is described furtherbelow with reference to the FIGS. 4 and 5. It should be noted that, inother embodiments, the memory elements 64 may include other types ofvolatile or nonvolatile memory. For example, the memory elements 64 mayinclude a resistive memory, such as a phase change memory ormagnetoresistive memory. In another example, the memory elements 64 mayinclude a capacitor, such as a stacked or trench capacitor. Some typesof memory elements 64 may include an access device, such as a transistoror a diode associated with each of the memory elements 64, or the memoryelements 64 may not include an access device, for instance in across-point array.

FIG. 4 illustrates a circuit 66 that models the operation of anarbitrarily selected memory element 64, which is disposed at theintersection of WL3 and BL0. This circuit 66 includes a capacitor 68, apre-drain resistor 70 (R_(PD)), a post-source resistor 72 (R_(PS)), anda ground 74. The resistors 70 and 72 model the other devices in seriesthe memory element 64 being sensed. The illustrated memory element 64includes a gate 76, a floating gate 78, a drain 80, and a source 82. Inthe circuit 66, the drain 80 and source 82 are disposed in seriesbetween the pre-drain resistor 70 and the post-source resistor 72. Thegate 76 is coupled to WL3. The pre-drain resistor 70, the drain 80, thesource 82, and the post-source resistor 72 are disposed in series on thebit-line BL0. The capacitor 68, which models the capacitance of thebit-line, has one plate coupled to ground 74 and another plate coupledto the bit-line BL0, in parallel with the memory elements 64.

Several of the components of the circuit 66 represent phenomenonaffecting the memory elements 64 during operation. The pre-drainresistor 70 generally represents the drain-to-bitline resistance of thememory elements 64 coupled to the bit-line above (i.e., up current from)WL3 when these memory elements 64 are turned on, (e.g., during a readoperation). Similarly, the post source resistor 72 generally correspondsto the source-to-ground resistance of the memory elements 64 coupled tothe bit-line below WL3 when these memory element 64 is selected. Thecircuit 66 models electrical phenomena associated with reading thememory elements 64 at the intersection of WL3 and BL0.

The operation of the memory elements 64 will now be briefly describedwith reference to FIGS. 4 and 5. FIG. 5 illustrates one potentialrelationship between the bit-line current (I_(Bit)), the word-linevoltage (V_(WL)), and the voltage of the floating gate 78 (V_(FG)). Asillustrated by FIG. 5, V_(FG) affects the response of the memory element64 to a given V_(WL). Decreasing the voltage of the floating gate shiftsthe I-V curve of the memory elements 64 to the right. That is, therelationship between the bit-line current and a word-line voltagedepends on the voltage of the floating gate 78. The memory elements 64may store and output data by exploiting this effect.

To write data to the memory elements 64, a charge corresponding to thedata may be stored on the floating gate 78. The charge of the floatinggate 78 may be modified by applying voltages to the source 82, drain 80,and/or gate 76 such that the resulting electric fields producephenomenon like Fowler-Northam tunneling and/or hot-electron injectionnear the floating gate 78. Initially, the memory elements 64 may beerased by manipulating the word-line voltage to drive electrons off ofthe floating gate 78. In some embodiments, an entire column or block ofmemory elements 64 may be erased generally simultaneously. Once thememory elements 64 are erased, the gate 76 voltage may be manipulated todrive a charge onto the floating gate 78 that is indicative of a datavalue. After the write operation ends, the stored charge may remain onthe floating gate 78 (i.e., the memory elements 64 may store data in anonvolatile fashion).

As illustrated by FIG. 5, the value stored by the memory element 64 maybe read by applying a voltage, V_(WL), to the gate 76 and measuring aresulting bit-line current, I_(Bit). Each of the I-V traces depicted byFIG. 5 correspond to a different charge stored on the floating gate,V_(FG), which should not be confused with the voltage that is applied tothe gate, V_(WL). The difference in floating gate 70 voltage, V_(FG),between each I-V trace is an arbitrarily selected scaling factor “x.”The illustrated I-V traces correspond to eight-different data valuesstored by the memory element 64, with a V_(FG) of 0x representing abinary data value of 000, a V_(FG) of 1x representing a binary datavalue of 001, and so on through V_(FG) of 7x, which represents a binarydata value of 111. Thus, by applying a voltage to the gate 76 andmeasuring the resulting bit-line current, the charge stored on thefloating gate 78 may be measured, and the stored data may be read.

The accuracy with which the bit-line current is sensed may affect theamount of data that a designer attempts to store in each memory element64. For example, in a system with a low sensitivity, a single bit may bestored on each memory element 64. In such a system, a floating gatevoltage V_(FG) of 0x may correspond to a value of 0, and a floating gatevoltage V_(FG) of −7x may correspond to a value of one. Thus, thedifference in floating gate voltages V_(FG) corresponding to differentdata values may be relatively large, and the resulting differences andbit-line currents for different data values may also be relativelylarge. As a result, even low-sensitivity sensing circuitry may discernthese large differences in bit-line current during a read operation. Incontrast, high-sensitivity sensing circuitry may facilitate storing moredata in each memory element 64. For instance, if the sensing circuitrycan distinguish between the eight different I-V traces depicted by FIG.5, then the memory elements 64 may store three bits. That is, each ofthe eight different charges stored on the floating gate 78 maycorrespond to a different three-bit value: 000, 001, 010, 011, 100, 101,110, or 111. Thus, circuitry that precisely measures the bit-linecurrent I_(BIT) may allow a designer to increase the amount of datastored in each memory element 64.

However, as mentioned above, a variety of effects may interfere withaccurate measurement of the bit-line current. For instance, the positionof the memory elements 64 along a bit-line may affect R_(PD) and R_(PS),which may affect the relationship between the word-line voltage V_(WL),and the bit-line current I_(BIT). To illustrate these effects, FIG. 6depicts noise on the bit-line while reading from the memory element 64.As illustrated, noise in the bit-line current I_(BIT) may cause thebit-line current I_(BIT) to fluctuate. Occasionally, the fluctuation maybe large enough to cause the bit-line current I_(BIT) to reach a levelthat corresponds with a different stored data value, which could causethe wrong value to be read from the memory elements 64. For instance, ifthe bit-line current is sensed at time 84, corresponding to anarbitrarily selected peak, a data value of 100 may be read rather thanthe correct data value of 011. Similarly, if the bit-line current issensed at time 86, corresponding to an arbitrarily selected localminimum, a data value of 010 may be read rather than a data value of011. Thus, noise on the bit-line may cause erroneous readings frommemory elements 64.

FIG. 7 depicts a quantizing circuit 16 that may tend to reduce thelikelihood of an erroneous reading. The illustrated quantizing circuit16 includes an analog-to-digital converter 88 and a digital filter 90coupled to each of the bit-lines 38, 40, 42, 44, and 46, respectively.That is, each bit-line 38, 40, 42, 44, and 46 may connect to a differentanalog-to-digital converter 88 and digital filter 90. The digitalfilters 90, in turn, may connect to an input/output bus 92, which mayconnect to a column decoder 18, a column address latch 20, and/orcontrol circuitry 28 (see FIG. 2).

In operation, the quantizing circuit 16 may digitize analog signals fromthe memory elements 64 in a manner that is relatively robust to noise.As explained below, the quantizing circuit 16 may do this by convertingthe analog signals into a bit-stream and digitally filteringhigh-frequency components from the bit-stream.

The analog-to-digital converter 88 may be a one-bit, analog-to-digitalconverter or a multi-bit, analog-to-digital converter. In the presentembodiment, an analog-to-digital converter 88 receives an analog signalfrom the memory element 64, e.g., a bit-line current I_(BIT) or abit-line voltage V_(BL), and outputs a bit-stream that corresponds withthe analog signal. The bit-stream may be a one-bit, serial signal with atime-averaged value that generally represents or corresponds to thetime-averaged value of the analog signal from the memory element 64.That is, the bit-stream may fluctuate between values of zero and one,but its average value, over a sufficiently large period of time, may beproportional to the average value of the analog signal from the memoryelement 64. In certain embodiments, the bit-stream from theanalog-to-digital converter 88 may be a pulse-density modulated (PDM)version of the analog signal. The analog-to-digital converter 88 maytransmit the bit-stream to the digital filter 90 on a bit-stream signalpath 94.

The digital filter 90 may remove high-frequency noise from thebit-stream. To this end, the digital filter 90 may be a low-pass filter,such as a counter, configured to average or integrate the bit-streamover a sensing time, i.e., the time period over which the memory element64 is read. As a result, the digital filter 90 may output a value thatis representative of both the average value of the bit-stream and theaverage value of the analog signal from the memory element 64. In someembodiments, the digital filter 90 is a counter, and the cut-offfrequency of the digital filter 90 may be selected by adjusting theduration of the sensing time. In the present embodiment, increasing thesensing time will lower the cutoff frequency. That is, the frequencyresponse of the digital filter 90 may be tuned by adjusting the periodof time over which the bit-stream is integrated and/or averaged beforeoutputting a final value. The frequency response of the digital filter90 is described further below with reference to FIG. 15. For multi-bitmemory elements 64, the output from the digital filter 90 may be amulti-bit binary signal, e.g., a digital word that is transmittedserially and/or in parallel.

Advantageously, in certain embodiments, the quantizing circuit 16 mayfacilitate the use of multi-bit memory elements 64. As described above,in traditional designs, the number of discrete data values that a memoryelement 64 stores may be limited by sense amps that react to noise. Incontrast, the quantizing circuit 16 may be less susceptible to noise,and, as a result, the memory elements 64 may be configured to storeadditional data. Without the high frequency noise, the intervals betweensignals representative of different data values may be made smaller, andthe number of data values stored by a given memory element 64 may beincreased. Thus, beneficially, the quantizing circuit 16 may sensememory elements 64 that store several bits of data, e.g., 2, 3, 4, 5, 6,7, 8, or more bits per memory element 64.

Although the quantizing circuit 16 may sample the signal from the memoryelement 64 over a longer period of time than conventional designs, theoverall speed of the memory device 12 may be improved. As compared to aconventional device, each read or write operation of the memory device12 may transfer more bits of data into or out of the memory element 64.As a result, while each read or write operation may take longer, moredata may be read or written during the operation, thereby improvingoverall performance. Further, in some memory devices 12, certainprocesses may be performed in parallel with a read or write operation,thereby further reducing the overall impact of the longer sensing time.For example, in some embodiments, the memory array 14 may be dividedinto banks that operate at least partially independently, so that, whiledata is being written or read from one bank, another bank can read orwrite data in parallel.

FIG. 8 illustrates details of one implementation of the quantizingcircuit 16. In this embodiment, the digital filter 90 is a counter, andthe analog-to-digital converter 88 is a first-order delta-sigmamodulator. The illustrated delta-sigma modulator 88 may include alatched comparator 96 (hereinafter the “comparator”), a capacitor 98,and a switch 100. In other embodiments, other types of digital filtersand analog-to-digital converters may be employed, such as thosedescribed below in reference to FIGS. 17 and 18.

As illustrated, an input of the counter 90 may connect to the bit-streamsignal path 94, which may connect to an output of the comparator 96. Theoutput of the comparator 96 may also connect to a gate of the switch 100by a feedback signal path 102. The output terminal (e.g., source ordrain) of the switch 100 may connect in series to one of the bit-lines38, 40, 42, 44, or 46, and the input terminal of the switch 100 mayconnect to a reference current source 104 (I_(Ref)). One plate of thecapacitor 98 may connect to one of the bit-lines 38, 40, 42, 44, or 46,and the other plate of the capacitor 98 may connect to ground.

The illustrated counter 90 counts the number of clock cycles that thebit-stream 94 is at a logic high value or logic low value during thesampling period. The counter may count up or count down, depending onthe embodiment. In some embodiments, the counter 90 may do both,counting up one for each clock cycle that the bit-stream has a logichigh value and down one for each clock cycle that the bit-stream has alogic low value. Output terminals (D0-D5) of the counter 90 may connectto the input/output bus 92 for transmitting the count. The counter 90may be configured to be reset to zero or some other value when a resetsignal is asserted. In some embodiments, the counter 90 may be a seriesconnection of D-flip flops, e.g., D-flip flops having SRAM or othermemory for storing an initial value and/or values to be written to thememory element 64.

In the illustrated embodiment, the clocked comparator 96 compares areference voltage (V_(Ref)) to the voltage of one of the bit-lines 38,40, 42, 44, or 46 (V_(BL)), which may be generally equal to the voltageof one plate of the capacitor 98. The comparator 96 may be clocked(e.g., falling and/or rising edge triggered), and the comparison may beperformed at regular intervals based on the clock signal, e.g., once perclock cycle. Additionally, the comparator 96 may latch, i.e., continueto output, values (V_(FB)) between comparisons. Thus, when the clocksignals the comparator 96 to perform a comparison, if V_(BL) is lessthan V_(Ref,) then the comparator 96 may latch its output to a logic lowvalue, as described below in reference to FIG. 9. Conversely, if V_(BL)is greater than V_(Ref), then the comparator 96 may latch a logic highvalue on its output, as described below in reference to FIG. 10. As aresult, the illustrated comparator 96 outputs a bit-stream thatindicates whether V_(BL) is larger than V_(Ref), where the indication isupdated once per clock cycle.

Advantageously, in some embodiments, the quantizing circuit 16 mayinclude a single comparator (e.g., not more than one) for each column ofmulti-level memory elements 64. In contrast, conventional senseamplifiers often include multiple comparators to read from a multi-bitmemory cell, thereby potentially increasing device complexity and cost.

The capacitor 98 may be formed by capacitive coupling of the bit-lines38, 40, 42, 44, and 46. In other designs, this type of capacitance isreferred to as parasitic capacitance because it often hinders theoperation of the device. However, in this embodiment, the capacitor 98may be used to integrate differences between currents on the bit-lines38, 40, 42, 44, or 46 and the reference current to form the bit-stream,as explained further below. In some embodiments, the capacitor 98 may besupplemented or replaced with an integrated capacitor that providesgreater capacitance than the “parasitic” bit-line capacitance.

The illustrated switch 100 selectively transmits current I_(Ref) fromthe reference current source 104. In various embodiments, the switch 100may be a PMOS transistor (as illustrated in FIGS. 8-10) or an NMOStransistor (as illustrated in FIG. 17) controlled by the V_(FB) signalon the feedback signal path 102.

The operation of the quantizing circuit 16 will now be described withreference to FIGS. 9-12. Specifically, FIGS. 9 and 10 depict currentflows in the quantizing circuit 16 when the comparator 96 is latched lowand high, respectively. FIG. 11 illustrates V_(BL), the bit-streamoutput from the comparator 96, and the corresponding increasing count ofthe counter 90 for a relatively small bit-line current. FIG. 12 depictsthe same voltages when measuring a medium sized bit-line current, andFIG. 13 depicts these voltages when measuring a relatively largebit-line current.

To measure the current through the memory element 64, the illustrateddelta-sigma modulator 88 exploits transient effects to generate abit-stream representative of the bit-line current I_(BIT). Specifically,the delta-sigma modulator 88 may repeatedly charge and discharge thecapacitor 98 with a current divider that subtracts the bit-line currentI_(BIT) from the reference current I_(REF). Consequently, a largecurrent through the memory element 64 may rapidly discharge thecapacitor 98, and a small current through the memory element 64 mayslowly discharge the capacitor 98.

To charge and discharge the capacitor 98, the delta-sigma modulator 88switches between two states: the state depicted by FIG. 9 (hereinafter“the charging state”) and the state depicted by FIG. 10 (hereinafter“the discharging state”). Each time the delta-sigma modulator 88 changesbetween these states, the bit-stream changes from a logic high value toa logic low value or vice versa. The proportion of time that thedelta-sigma modulator 88 is in the state illustrated by either FIG. 9 orFIG. 10 may be proportional to the size of the bit-line current I_(BIT)through the memory element 64. The larger the bit-line current I_(BIT),the more time that the delta-sigma modulator 88 is in the stateillustrated by FIG. 9, rather than the state illustrated by FIG. 10, andthe more time that the bit-stream has a logic low value.

Starting with the charging state (FIG. 9), the capacitor 98 mayinitially accumulate a charge. To this end, the output of the comparator96 is latched to logic low, which, as mentioned above, may occur whenV_(BL) is less than V_(Ref). The logic low may be conveyed to switch 100by the feedback signal path 102, and the switch 100 may close, therebyconducting the reference current I_(Ref) through one of the bit-lines38, 40, 42, 44, or 46, as indicated by the larger arrows in FIG. 9. Aportion of the electrons flowing through the reference current source104 may be stored by the capacitor 98, as indicated by thesmaller-horizontal arrows, and the remainder may be conducted throughthe memory element 64, i.e., the bit-line current I_(Bit), as indicatedby the smaller vertical arrows. Thus, the capacitor 98 may accumulate acharge, and V_(BL) may increase.

The comparator 96 and the reference current source 104 may cooperate tocharge the capacitor 98 for a discrete number of clock cycles. That is,when the delta-sigma modulator 88 enters the charging state, thedelta-sigma modulator 88 may remain in this state for an integer numberof clock cycles. In the illustrated embodiment, the comparator 96, theoutput of which is latched, changes state no more than once per clockcycle, so the switch 100, which is controlled by the output of thecomparator 96, V_(FB), conducts current for a discrete number of clockcycles. As a result, the reference current source 104 conducts currentI_(Ref) through the bit-line and into the capacitor 98 for an integernumber of clock cycles.

After each clock cycle of charging the capacitor 98, the delta-sigmamodulator 88 may transition from the charging state to the dischargingstate, which is illustrated by FIG. 10, depending on the relative valuesof V_(BL) and V_(Ref). Once per clock cycle (or at some otherappropriate interval, such as twice per clock cycle), the comparator 96may compare the voltage of the capacitor V_(BL) to the reference voltageV_(Ref). If the capacitor 98 has been charged to the point that V_(BL)is greater than V_(Ref), then the output of the comparator 96 maytransition to logic high, as illustrated in FIG. 10. The logic highsignal may be conveyed to the switch 100 by the feedback signal path102, thereby opening the switch 100. As a result, the reference currentsource 104 may cease flowing current through the memory element 64 andinto the capacitor 98, and the capacitor 98 may begin to dischargethrough the memory element 64.

In the present embodiment, the delta-sigma modulator 88 discharges thecapacitor 98 for a discrete number of clock intervals. After each clockcycle of discharging the capacitor 98, the delta-sigma modulator 88compares V_(BL) to V_(Ref). If V_(BL) is still greater than V_(Ref),then the comparator 96 may continue to output a logic high signal, i.e.,V_(FB)=1, and the switch 100 remains open. On the other hand, if enoughcurrent has flowed out of the capacitor 98 that V_(BL) is less thanV_(Ref), then the comparator 96 may output a logic low signal, i.e.,V_(FB)=0, and the switch 100 may close, thereby transitioning thedelta-sigma modulator 88 back to the charging state and initiating a newcycle.

The counter 90 may count the number of clock cycles that the delta-sigmamodulator 88 is in either the charging state or the discharging state bymonitoring the bit-stream signal path 94. The bit-stream signal path 94may transition back and forth between logic high and logic low with theoutput of the comparator 96, V_(FB), and the counter 90 may incrementand/or decrement a count once per clock cycle (or other appropriateinterval) based on whether the bit-stream is logic high or logic low.After the sensing time has passed, the counter 90 may output a signalindicative of the count on output terminals D0-D5. As explained below,the count may correspond, e.g., proportionally, to the bit-line current,I_(Bit).

FIGS. 11-13 illustrate voltages V_(FB) and V_(BL) in the quantizingcircuit 16 when reading a memory element 64. Specifically, FIG. 11illustrates a low-current case, in which the value stored by the memoryelement 64 corresponds to a relatively low bit-line current. Similarly,FIG. 12 illustrates a medium-current case, and FIG. 13 illustrates ahigh-current case. In each of these figures, the ordinate of the lowertrace represents the voltage of the bit-stream signal path 94, V_(FB),and the ordinate of the upper trace illustrates the bit-line voltage,V_(BL). The abscissa in each of the traces represents time, with thelower trace synchronized with the upper trace, and the duration of thetime axes is one sensing time 106.

As illustrated by FIG. 11, the counter 90 is initially set to zero (orsome other appropriate value) by asserting a reset signal. In someembodiments, the delta-sigma modulator 88 may undergo a number ofstart-up cycles to reach steady-state operation before initiating thesensing time and resetting the counter 90. At the beginning of theillustrated read operation, the delta-sigma modulator 88 is in thecharging state, which charges the capacitor 98 and increases V_(BL), asindicated by dimension arrow 108. At the beginning of the next clockcycle, the comparator 96 compares the bit-line voltage to the referencevoltage and determines that the bit-line voltage is greater than thereference voltage. As a result, the bit-stream signal path 94 (V_(FB))transitions to a logic high voltage, and the delta-sigma modulator 88transitions to the discharging state. Additionally, the counter 90increments the count by one to account for one clock cycle of thebit-stream signal 94 holding a logic low value. Next, the charge storedon the capacitor 98 drains out through the memory element 64, and thebit-line voltage drops until the comparator 96 detects that V_(BL) isless than V_(Ref), at which point the cycle repeats. The cycle has aperiod 112, which may be divided into a charging portion 114 and adischarging portion 116. Once during each cycle in the sensing time 106,the count stored in the counter 90 may increase by one. At the end ofthe sensing time 106, the counter 90 may output the total count.

A comparison of FIG. 11 to FIGS. 12 and 13 illustrates why the countcorrelates with the bit-line current. In FIG. 13, the high-current case,the stored charge drains from the capacitor 98 quickly, relative to theother cases, because the bit-line current I_(BIT) is large and, as aresult, the delta-sigma modulator 88 spends more time in the chargingstate than the discharging state. As a result, the bit-stream has alogic low value for a large portion of the sensing time 106, therebyincreasing the count.

The capacitance of the capacitor 98 may be selected with both the clockfrequency and the range of expected bit-line currents in mind. Forexample, the capacitor 98 may be large enough that the capacitor 98 doesnot fully discharge or saturate when the bit-line current I_(BIT) iseither at its lowest expected value or at its highest expected value.That is, in some embodiments, the capacitor 98 generally remains in atransient state while reading the memory element 64. Similarly, thefrequency at which the comparator 96 is clocked may affect the design ofthe capacitor 98. A relatively high frequency clock signal may leave thecapacitor 98 with relatively little time to discharge or saturatebetween clock cycles, thereby leading a designer to choose a smallercapacitor 98.

Similarly, the size of the reference current may be selected with therange of expected bit-line currents in mind. Specifically, in certainembodiments, the reference current is less than the largest expectedbit-line current I_(BIT), so that, in the case of maximum bit-linecurrent I_(BIT,) the capacitor 98 can draw charge from the referencecurrent while the rest of the reference current flows through the memoryelement 64.

FIG. 14 illustrates the relationship between the bit-line currentI_(BIT) and the count for the presently discussed embodiment. Asillustrated by FIG. 14, the count is generally proportional to thebit-line current I_(BIT). This relationship is described by thefollowing equation (Equation 1), in which N_(ST) represents the numberof clock cycles during the sensing time:

I _(Bit) /I _(Ref)=Count/N _(ST)

Thus, in the illustrated embodiment, the count is indicative of thebit-line current I_(BIT), which is indicative of the value stored by thememory element 64.

Advantageously, the quantizing circuit 16 may categorize the bit-linecurrent I_(BIT) as falling into one of a large number of categories,each of which is represented by an increment of the count. That is, thequantizing circuit 16 may resolve small differences in the bit-linecurrent I_(BIT). The resolution of the quantizing circuit 16 may becharacterized by the following equation (Equation 2), in which I_(MR)represents the smallest resolvable difference in bit-line currentI_(BIT), i.e., the resolution of the quantizing circuit 16:

I _(MR) =I _(Ref) /N _(ST)

Thus, the resolution of the quantizing circuit 16 may be increased byincreasing the sensing time or the clock frequency or by decreasingI_(Ref), which may limit the maximum cell current since I_(MR) is lessthan I_(Ref).

The resolution of the quantizing circuit 16 may facilitate storingmultiple bits in the memory element 64 or detecting multiple levels oflight intensity in an image sensor element. For example, if thequantizing circuit 16 is configured to categorize the bit-line currentI_(BIT) into one of four different levels, then the memory element 64may store two-bits of data or, if the quantizing circuit 16 isconfigured to categorize the bit-line current I_(BIT) into one of eightdifferent current levels, then the memory element 64 may storethree-bits of data. For the present embodiment, the number of bitsstored by the memory element 64 may be characterized by the followingequation (Equation 3), in which N_(B) represents the number of bitsstored by a memory element 64 and I_(Range) represents the range ofprogrammable bit-line currents through the memory element 64:

N _(B)=log(I _(Range) /I _(MR))/log 2

In short, in the present embodiment, greater resolution translates intohigher density data storage for a given memory element 64.

FIG. 15 is a graph that illustrates one way in which the counter 90 maybe configured to further reduce the effects of noise. In FIG. 15, theabscissa represents the count, and the ordinate represents the output ofthe quantizing circuit 16. In the present embodiment, thethree-least-significant digits of the count are disregarded aspotentially corrupted by noise. That is, D0-D2 (FIG. 8) either do notconnect to the input/output bus 92 or are not interpreted as conveyingdata that is stored by the memory element 64. As a result, a range ofcounter values may represent a single data value stored by the memoryelement 64. For example, in the present embodiment, count values rangingfrom 00 1000 to 00 1111 are construed as representing a data value of001. Representing data in this manner may further reduce the effects ofnoise because, even if noise affects the count, in many embodiments, itwould have to affect the count in a consistent manner over a substantialportion of the sensing time to affect the more significant digits of thecount. That is, disregarding less significant digits may lower thecutoff frequency of the counter 90. In other embodiments, fewer, more,or no digits may be truncated from the count as potentially representingnoise.

Truncating less significant digits may introduce a rounding error, or adownward bias, in the output. This effect may be mitigated by presettingthe counter 90 in a manner that accounts for this bias. The counter 90may be present either before reading from the memory element 64 orbefore writing to the memory element 64. In some embodiments, the presetvalue may be one-half of the size of the range of counter values thatrepresent a single output value. In other words, if m digits aretruncated from the output, then the counter 90 may be preset to one-halfof 2^(m) before reading from a memory element 64 or before writing tothe memory element 64. In some embodiments, the memory 91 may store thispreset value.

Delta-sigma modulators may be formed with a variety of circuittopologies. A broad array of these topologies is illustrated by FIG. 16,which is a block diagram of an example of a first-order delta-sigmamodulator 120. As described below, the embodiment of FIG. 16 is genericto the delta-sigma modulator 88 illustrated in FIG. 8. The illustrateddelta-sigma modulator 120 includes an adder 122, an integrator 124, ananalog-to-digital converter (A/D) 126, and a digital-to-analog converter(D/A) 128. The illustrated adder 122 receives an analog input signal 130and a feedback signal 132 from the digital-to-analog converter 128. Theillustrated adder 122 outputs a delta signal 134 to an input of theintegrator 124, which outputs a sigma signal 136 to an input of theanalog-to-digital converter 126. The analog-to-digital converter 126also receives a reference signal 138. The analog-to-digital converter126 outputs a digital output signal 140, which is received by an inputto the digital-to-analog converter 128.

FIG. 17 is a block diagram of an example of a one-bit delta-sigmamodulator 142, which is an embodiment of the delta-sigma modulator 120illustrated by FIG. 16, and which generic to the delta-sigma modulator88 illustrated by FIG. 8. In this example, the integrator 124 is acapacitor and the analog-to-digital converter 126 is a comparator. Thereference signal 138 is a voltage V_(REF), and the digital-to-analogconverter 128 includes a switch 144 and a reference current source 146.

In operation, the illustrated delta-sigma modulator 142 measures theanalog input signal 130 by integrating a difference between the analoginput signal and the feedback signal 132 and exercising feedback controlover this integrated difference. The greater the difference, thestronger or the more frequent the feedback signal 132. The analogsignals may be voltage signal or current signals. Examples thatintegrate a difference in analog voltage signals are described below.Previously, an example of an analog current signal was described inreference to FIG. 8. In this embodiment, the difference between thebit-line current I_(Bit) and the reference current I_(Ref) is integratedby the voltage of the capacitor 98, and the comparator 96 controls thisvoltage by outputting feedback on the feedback signal path 102. Incertain embodiments, if the strength of the feedback signal is heldrelatively constant when it is applied, the proportion of time that thefeedback signal is applied is indicative of the analog input signal 130.Thus, consistently applying a feedback signal of the same strength mayimprove the correlation between the digital output 140 and the analoginput 130, thereby potentially improving the accuracy of the quantizingcircuit 16.

The following figures illustrate embodiments of the adder 122 and thedigital-to-analog converter 128 that may apply the feedback signal 132in a relatively consistent manner. FIGS. 18-22 illustrate addersconfigured to add the analog input signal 130 and the feedback signal132 (or its inverse), where the analog input signal 130 and/or thefeedback signal 132 is a voltage, and FIGS. 23-29 illustrate examples ofthe digital-to-analog converter 128. Some of the following embodimentsare believed to improve the precision of the delta-sigma modulators 120and/or 142 and/or lower the power consumption of these componentsrelative to conventional designs, as described below.

FIG. 18 illustrates a first embodiment of the adder 122 of FIGS. 16 and17. This embodiment is designated with reference number 148. Theillustrated adder 148 includes voltage-to-current converters 150 and152. In the present embodiment, the input terminal of thevoltage-to-current converter 150 is connected to an analog input signalV_(IN-Analog), and the input terminal of the voltage-to-currentconverter 152 is connected to a voltage feedback signal V_(Feedback).Details of examples of voltage-to-current converters are described belowwith reference to the FIGS. 19 and 20. In operation, thevoltage-to-current converter 150 may convert V_(IN-Analog) to a currentI_(in), and the voltage-to-current converter 152 may convertV_(Feedback) to a current I_(f). The output terminals of thevoltage-to-current converters 150 and 152 may be coupled to one anotherand an adder output I_(in)+I_(f). Thus, the adder 148 may be configuredto convert voltage signals to currents and combine those currents.

In other embodiments, the adder 148 may convert only one voltage signalto a current and combine that current with another signal that isalready in the form of a current. Also, in some embodiments, the currentoutputs from the voltage-to-current converters I_(in) and I_(f) may havedifferent signs, and the adder 148 may output a difference of thesecurrents, i.e., the adder 148 may also subtract.

FIG. 19 illustrates an example of a voltage-to-current converter 154,which may embody one or both of the voltage-to-current converters 150and 152 illustrated in FIG. 18. In this example, the voltage-to-currentconverter 154 includes an NMOS transistor 156 and a resistor 158. Theresistor 158 may be disposed in series between ground 74 and the sourceof the transistor 156, and an input voltage V_(IN), such asV_(IN-Analog) or V_(feedback) in FIG. 18, may connect to the gate of thetransistor 156. In the present embodiment, the transistor 156 has a gatewidth sized such that a voltage difference between the gate and thesource V_(GS) is near the threshold voltage of the transistor V_(THN).Consequently, in certain embodiments, the transistor 156 operates in thelinear region of its I-V curve. As a result, in some embodiments, theoperation of the voltage-to-current converter 154 may be characterizedby the following equation (Equation 4), in which I corresponds to theoutput current signal and R corresponds to the resistance of theresistor 158:

I=(V _(IN) −V _(THN))/R

FIG. 20 illustrates another example of a voltage-to-current converter160, which may embody one or both of the voltage-to-current converters150 and 152 illustrated in FIG. 18. The voltage-to-current converter 160may include a PMOS transistor 162 (in contrast to the NMOS transistor ofthe previous embodiment) and a resistor 164, which may be disposed inseries between the transistor 162 and a voltage source V_(DD). Thetransistor 162 may have a relatively wide gate that is sized such thatits operation may be described by the following equation (Equation 5),wherein V_(THP) is the threshold voltage of the transistor 162, I is theoutput current signal, and R is the resistance of the resistor 164:

I=(V _(DD) −V _(THP) −V _(IN))/R

FIGS. 21 and 22 illustrate examples of the adder 148 (FIG. 18) that mayemploy the voltage-to-current converters 154 and 160 (FIGS. 19 and 20).In FIG. 21, the adder 151 sources the current I_(IN)+I_(F). The adder151 may include two voltage-to-current converters 153 and 155, which maycorrespond to the voltage-to-current converter 154 of FIG. 19, and twocurrent mirrors 157 and 159. In FIG. 22, the adder 161 sinks the currentI_(IN)+I_(F). The adder 161 may include voltage-to-current converters163 and 165, which may correspond to the voltage-to-current converter160 of FIG. 20, and two current mirrors 167 and 169.

FIG. 23 illustrates another example of an adder 166, which may embodythe adder 122 illustrated in FIGS. 16 and 17. In this embodiment, theadder 166 includes a pair of voltage-to-current converters 168 and 170that are configured to convert a voltage signal into a current signalthat may span two directions of current flow. That is, the output of thevoltage-to-current converters ±I_(IN) and ±I_(f) may flow in bothdirections, depending on the input voltage signal ±V_(IN) or ±V_(f).Consequently, the output ±I_(IN) ±I_(f) may also range over twodirections of current flow.

FIG. 24 illustrates a voltage-to-current converter 172, which may embodyone or both of the voltage-to-current converters 168 and 170 illustratedin FIG. 23. The present voltage-to-current converter 172 may include aforward-current converter 174 and a reverse-current converter 176.

The forward-current converter 174 may include two-PMOS transistors 178and 180, an NMOS transistor 182, and a resistor 184. The resistor 184may be disposed in series between the transistor 182 and ground 74. Thetransistor 178 may be disposed in series between the voltage sourceV_(DD) and the transistor 182. The transistors 178 and 180 may share agate signal 188, which may be coupled to a node in series between thetransistors 178 and 182. The transistor 180 may be disposed in seriesbetween the voltage source V_(DD) and an output signal path 190, whichcarries an output current signal ±I. The gate of the transistor 182 maybe connected to an input signal path 192, which carries an input voltagesignal ±V_(IN). The transistors 178, 180, and 182 may have relativelywide gate widths, so that, during normal operation, they operate in thelinear region of their I-V curves.

Similarly, the reverse-current converter 176 may include two-NMOStransistors 194 and 196, a PMOS transistor 198, and a resistor 200. Theresistor 200 may be disposed in series between the voltage source V_(DD)and the transistor 198. The transistor 194 may be disposed in seriesbetween ground 74 and the transistor 198. The transistors 194 and 196may share a gate signal 202, which may be coupled to a node in seriesbetween the transistors 198 and 194. The transistor 196 may be disposedin series between ground 74 and the output signal path 190. The gate oftransistor 198 may be connected to the input signal path 192. Thetransistors 198, 194, and 196 may have relatively wide gate widths, sothat, during normal operation, they operate in the linear region oftheir I-V curves.

In operation, either the forward-current converter 174 may act as acurrent source or the reverse-current converter 176 may act as a currentsink, depending on the input voltage signal ±V_(IN). If the inputvoltage signal ±V_(IN) is positive, then the transistor 198 may turnoff, and the transistor 182 may turn on. As a result, in someembodiments, a current I₁ may flow through the transistor 182, and acurrent I₂ may cease flowing through transistor 198. Because thetransistors operate in the linear region of their I-V curve, themagnitude of the currents I₁ and I₂ may be generally proportional to themagnitude of the input voltage signal ±V_(IN). The transistors 178 and180 may mirror the current I₁ on the output signal path 190, and thetransistors 194 and 196 may mirror the current I₂ on the output signalpath 190.

Advantageously, the current mirrors formed by the transistors 178, 180,194, and 196 may form the output current ±I in a manner that isrelatively robust to loads applied to the output signal path 190. As aresult, the adder 122 (FIGS. 16 and 17) may receive signals thataccurately convey information about the analog input or the state of thedelta-sigma modulation circuit 120 or 142, such as the magnitude of aninput voltage or a feedback voltage. This is believed to improve theaccuracy of the adder.

FIG. 25 illustrates an example of a switch 204, which may be employed inplace of the current switch 100 in the delta-sigma modulation circuit 88illustrated by FIG. 8. The illustrated switch 204 routes the referencecurrent I_(Ref) either to ground 74 or to a bit-line 208 based on thefeedback signal V_(feedback) from the comparator 96 (FIG. 8) on thefeedback signal path 102. In certain embodiments, the switch 204converts a digital signal from the analog-to-digital converter 126,i.e., a digital feedback signal, to an analog signal, i.e., thereference current. The switch 204 may include two PMOS transistors 210and 212, the gates of which may be controlled by V_(feedback) or itscomplement.

Advantageously, the switch 204 may supply a relatively uniform referencecurrent I_(Ref) because of the reference current I_(Ref) flows generallyconstantly, rather than turning on and off. As a result, transienteffects, such as parasitic induction and capacitance, that may distortthe reference current I_(Ref) in other designs may be avoided. This isbelieved to improve the accuracy of the delta-sigma modulator 88 (FIG.8) because the bit-line current I_(Bit) is measured against, e.g.,subtracted from, a relatively constant quantity.

FIG. 26 illustrates another example of a switch 214, which may beemployed in place of the current switch 100 in the delta-signalmodulation circuit 88 illustrated by FIG. 8. The switch 214 may includePMOS transistors 216 and 218 and a capacitor 220. In the presentembodiment, one plate of the capacitor 220 is coupled to ground 74 andthe other plate is coupled to both the reference voltage V_(Ref) (whichis also coupled to the comparator 96 in FIG. 8) and one terminal of thetransistor 216. The other terminal of the transistor 216 may beconnected to the reference current source 104 along with one terminal ofthe transistor 218. The gates of the transistors 216 and 218 may beconnected to the feedback signal path 102 to receive V_(feedback).

In operation, the capacitor 220 may supplement the reference currentsource 104 during transient periods with an elevated load. When thetransistors 216 and 218 turn on, and the reference current I_(Ref)begins to flow through the bit-line 208, the reference current source104 may be unable to supply a uniform current due to parasitic,transient effects. To counteract these transient effects, the referencevoltage may maintain a charge on the capacitor 220, which may supplycurrent to the bit-line 208 and the current from the reference currentsource 104. In other words, the capacitor 220 may store and releaseenergy to average out the load seen by the reference current source 104.As a result, the reference current I_(Ref) may be more uniform, which isbelieved to improve the accuracy of the delta-sigma modulation circuit88 (FIG. 8). In certain embodiments, the switch 214 converts a digitalsignal from the analog-to-digital converter 126 to an analog signal.

FIG. 27 illustrates another example of a switch 223, which may beemployed in place of the current switch 100 in the delta-signalmodulation circuit 88 illustrated by FIG. 8. The illustrated switch 223includes a PMOS transistor 224 and a capacitor 225. One terminal of thetransistor 224 may be connected to one plate of the capacitor 225 andthe reference current source 104. The other terminal of the transistor224 may be connected to the bit-line 208, and the gate of the transistor224 may be connected to the feedback signal path 102 to receiveV_(feedback) from the comparator 96 (FIG. 8). One plate of the capacitor225 may also be connected to ground.

In operation, the capacitor 225 may deliver supplemental power to thereference current source 104. When the transistor 224 turns on or off,the load on the reference current source 104 may spike, therebypotentially disrupting the reference current I_(Ref). To counteract thiseffect, the capacitor 225 may store charge from the reference currentI_(Ref) when the transistor 224 is turned off, and a release that chargethrough the bit-line 208 when the transistor 224 turns on. In certainembodiments, the switch 223 converts a digital signal from theanalog-to-digital converter 126 to an analog signal. Advantageously, theswitch 223 may provide a relatively stable reference current I_(Ref).Additionally, in the present embodiment, the switch 223 consumesrelatively little power because the reference current I_(Ref) does notflow to ground 74 when it is not flowing through the bit-line 208.

FIG. 28 illustrates an example of a reference current source 227, whichmay embody the reference current source 104 illustrated in theembodiments of FIGS. 8-10 and 23-24. As described below, the referencecurrent source 227 may cooperate with the switch 100 to converts adigital signal from the analog-to-digital converter 126 to an analogsignal. The illustrated reference current source 227 includes areference device 228 and a current mirror 229 with PMOS transistors 230and 232. The reference device 228 may include a reference resistor, areference memory element, or any other device configured to conduct arelatively uniform current. The reference device 228 may be disposed inseries between ground 74 and a terminal of the transistor 230. The otherterminal of the transistor 230 may be connected to the voltage sourceV_(DD). The gates of the transistors 230 and 232 may be connected to oneanother and to a node disposed in series between the transistors 230 andthe reference device 228. The terminals of the transistor 232 may beconnected to the voltage source V_(DD) and an output signal path 236through which the reference current I_(Ref) flows.

In operation, the current mirror 229 may copy the current flowingthrough the reference device 228 to the output signal path 236. Thecurrent mirror 229 may do this while keeping the reference currentI_(Ref) relatively constant regardless of heavy loads from the bit-line,e.g., immediately after the current switch 100 turns on (FIG. 8).Advantageously, this is believed to improve the accuracy of thedelta-sigma modulation circuit 88 (FIG. 8) relative to conventionaldesigns.

FIG. 29 illustrates another example of a reference current source 238,which may embody the reference current source 104 illustrated in theembodiments of FIGS. 8-10 and 23-24. As described below, the referencecurrent source 238 may cooperate with the switch 100 to converts adigital signal from the analog-to-digital converter 126 to an analogsignal. The illustrated reference current source 238 includes thefeatures of the reference current source 224 illustrated in FIG. 27.However, the reference current source 238 may also include additionalcomponents to form a cascade current mirror 240. The illustrated cascadecurrent mirror 240 includes the current mirror 228 and a second currentmirror 242. The second current mirror 242 may include to PMOStransistors 244 and 246. Advantageously, in systems employing thecascade current mirror 240 the reference current I_(Ref) may track thecurrent through the reference device 226 more accurately than systemsemploying a single current mirror.

FIG. 30 illustrates another example of a reference current source 248,which may embody the reference current source 104 illustrated in theembodiments of FIGS. 8-10 and 23-24. As described below, the referencecurrent source 248 may cooperate with the switch 100 to converts adigital signal from the analog-to-digital converter 126 to an analogsignal. The illustrated reference current source 248 includes a resistor250 and the switch 204, which was previously described with reference toFIG. 25. The resistor 250 may be disposed in series between the voltagesource V_(DD) and the switch 204.

In operation, the reference current source 248 may output a referencecurrent I_(Ref) that is generally proportional to the voltage betweenthe voltage source V_(DD) and the voltage of the bit-line V_(BL). Incertain embodiments, the feedback signal V_(feedback) from thecomparator 96 (FIG. 8) keeps the voltage of the bit-line V_(BL)approximately equal to the reference voltage V_(Ref). Although thevoltage of the bit-line V_(BL) fluctuates relative to the referencevoltage V_(Ref) (as illustrated by 11-13), this fluctuation may berelatively small as a percentage of the reference voltage V_(Ref). As aresult, for certain purposes, the voltage of the bit-line V_(BL) may beconsidered approximately equal to the reference voltage V_(Ref).Consequently, in the present embodiment, the reference current may bedescribed by the following equation (Equation 6), in which R is theresistance of the resistor 250:

I _(Ref)=(V _(DD) −V _(Ref))/R

FIG. 31 illustrates another example of a reference current source 252,which may embody the reference current source 104 illustrated in theembodiments of FIGS. 8-10 and 25-26. As described below, the referencecurrent source 252 may cooperate with the switch 100 to converts adigital feedback signal from the analog-to-digital converter 126 to ananalog signal. In the present embodiment, the reference current source252 includes two PMOS transistors 254 and 256 and a capacitor 258. Oneplate of the illustrated capacitor 258 is connected to a terminal ofeach of the transistors 254 and 256, and the other plate of thecapacitor 258 is connected to ground 74. The other terminal of thetransistor 254 is connected to voltage source V_(DD), and the otherterminal of the transistor 256 is connected to a reference currentsignal path 260, which carries the reference current I_(Ref). The gatesof the transistors 254 and 256 may be connected to complementary clocksignals. As a result, in the present embodiment, the transistors 254 and256 are neither both on at the same time nor both off at the same time.

In operation, the capacitor 258 may output charge at a generallyrepeatable rate. In the present embodiment, the capacitor 258 is chargedby a current from the voltage source V_(DD) when the clock signal islogic high. When the clock signal transitions to logic low, thecapacitor 258 discharges through the transistor 256. In this embodiment,assuming that V_(Ref) approximates V_(BL), the operation of thereference current source 252 may be described by the following equation(Equation 7), in which C₄ represents the capacitance of the capacitor258 and I_(Avg) represents the average reference current I_(Ref):

I _(Avg) =C ₄*(V _(DD) −V _(Ref))

Advantageously, the reference current source 252 may deliver a uniformreference current, on average, while consuming relatively little power.As indicated by Equation 7, in certain embodiments, the averagereference current I_(Avg) is a function of generally constantparameters, C₄, V_(DD), and V_(Ref). As a result, the average referencecurrent I_(Avg) may be generally the same when reading from, and writingto, the memory element 64. In certain embodiments, this increases thelikelihood that the value read from the memory element 64 accuratelyreflects the value written to the memory element 64. Further, becausethe capacitor 258 conserves charge and generates relatively little heatcompared to a resistor, the reference current source 252 may consumeless power than other designs.

FIG. 32 depicts an exemplary processor-based system 310 that includesthe memory device 12. Alternatively or additionally, the system 310 mayinclude the imaging device 13. The system 310 may be any of a variety oftypes such as a computer, pager, cellular phone, personal organizer,control circuit, etc. In a typical processor-based system, one or moreprocessors 312, such as a microprocessor, control the processing ofsystem functions and requests in the system 310. The processor 312 andother subcomponents of the system 310 may include quantizing circuits,such as those discussed above.

The system 310 typically includes a power supply 314. For instance, ifthe system 310 is a portable system, the power supply 314 mayadvantageously include a fuel cell, permanent batteries, replaceablebatteries, and/or rechargeable batteries. The power supply 314 may alsoinclude an AC adapter, so the system 310 may be plugged into a walloutlet, for instance. The power supply 314 may also include a DC adaptersuch that the system 310 may be plugged into a vehicle cigarettelighter, for instance.

Various other devices may be coupled to the processor 312 depending onthe functions that the system 310 performs. For instance, a userinterface 316 may be coupled to the processor 312. The user interface316 may include buttons, switches, a keyboard, a light pen, a mouse, adigitizer and stylus, and/or a voice recognition system, for instance. Adisplay 318 may also be coupled to the processor 312. The display 318may include an LCD, an SED display, a CRT display, a DLP display, aplasma display, an OLED display, LEDs, and/or an audio display, forexample. Furthermore, an RF sub-system/baseband processor 320 may alsobe coupled to the processor 312. The RF sub-system/baseband processor320 may include an antenna that is coupled to an RF receiver and to anRF transmitter (not shown). One or more communication ports 322 may alsobe coupled to the processor 312. The communication port 322 may beadapted to be coupled to one or more peripheral devices 324 such as amodem, a printer, a computer, or to a network, such as a local areanetwork, remote area network, intranet, or the Internet, for instance.

The processor 312 generally controls the system 310 by implementingsoftware programs stored in the memory. The memory is operably coupledto the processor 312 to store and facilitate execution of variousprograms. For instance, the processor 312 may be coupled to the volatilememory 326 which may include Dynamic Random Access Memory (DRAM) and/orStatic Random Access Memory (SRAM). The volatile memory 326 is typicallylarge so that it can store dynamically loaded applications and data. Asdescribed further below, the volatile memory 326 may be configured inaccordance with embodiments of the present invention.

The processor 312 may also be coupled to the memory device 12. Thememory device 12 may include a read-only memory (ROM), such as an EPROM,and/or flash memory to be used in conjunction with the volatile memory326. The size of the ROM is typically selected to be just large enoughto store any necessary operating system, application programs, and fixeddata. Additionally, the non-volatile memory 328 may include a highcapacity memory such as a tape or disk drive memory.

The memory device 10 and volatile memory 326 may store various types ofsoftware, such as an operating system or office productivity suiteincluding a word processing application, a spreadsheet application, anemail application, and/or a database application.

While the invention may be susceptible to various modifications andalternative forms, specific embodiments have been shown by way ofexample in the drawings and have been described in detail herein.However, it should be understood that the invention is not intended tobe limited to the particular forms disclosed. Rather, the invention isto cover all modifications, equivalents, and alternatives falling withinthe spirit and scope of the invention as defined by the followingappended claims.

What is claimed is:
 1. A method of sensing a memory element, the methodcomprising: integrating a difference between an analog input signal anda feedback signal to produce an integrated difference; and exercisingfeedback control over the integrated difference.
 2. The method of claim1, wherein integrating a difference comprises integrating a differencebetween an analog current signal and a feedback current signal toproduce an integrated current signal.
 3. The method of claim 1, whereinintegrating a difference comprises integrating a difference between anelectrical conductor current and a reference current by a voltage of acapacitor.
 4. The method of claim 3, wherein integrating the differencecomprises integrating a difference between a bit-line current and areference current by a voltage of a capacitor.
 5. The method of claim 4,wherein exercising feedback control comprises controlling the voltage ofthe capacitor.
 6. The method of claim 5, wherein controlling the voltagecomprises controlling the voltage using a comparator configured tooutput the feedback signal.
 7. The method of claim 1, wherein the analoginput signal is related to a value stored in the memory element.
 8. Adevice, comprising: a combination circuit configured to combine ananalog input signal with an analog feedback signal to produce a deltasignal; an integrator configured to receive and integrate the deltasignal to produce a sigma signal; and an analog-to-digital converterconfigured to receive the sigma signal and compare the sigma signal witha reference signal to produce a digital output signal.
 9. The device ofclaim 8, comprising a digital-to-analog converter configured to convertthe digital output signal to the analog feedback signal.
 10. The deviceof claim 9, wherein the digital-to-analog converter comprises a currentsource and a switch.
 11. The device of claim 8, wherein the integratorcomprises a capacitor.
 12. The device of claim 8, wherein theanalog-to-digital converter comprises a comparator.
 13. The device ofclaim 8, wherein the combination circuit comprises a first voltage tocurrent converter configured to convert the analog input signal to ananalog input current.
 14. The device of claim 13, wherein thecombination circuit comprises a second voltage to current converterconfigured to convert the analog feedback signal to an analog feedbackcurrent.
 15. The device of claim 13, wherein the first voltage tocurrent converter comprises a transistor in series with a resistor. 16.The device of claim 8, comprising an internal data storage locationcoupled to the combination circuit, wherein the analog input signal isrelated to a value stored in the data storage location.
 17. A device,comprising: a combination circuit configured to combine an analog inputsignal related to a value stored in a data storage location with ananalog feedback signal to produce a delta signal; an integratorconfigured to receive and integrate the delta signal to produce a sigmasignal; an analog-to-digital converter configured to receive the sigmasignal and compare the sigma signal with a reference signal to produce adigital output signal; and a digital-to-analog converter configured toconvert the digital output signal to the analog feedback signal.
 18. Thedevice of claim 17, wherein the digital-to-analog converter comprises aswitch configured to selectively route a reference current to ground orto conductor coupled to the data storage location based on the digitaloutput signal to generate the analog feedback signal.
 19. The device ofclaim 17, wherein the digital-to-analog converter comprises a switchcoupled to a capacitor, wherein the switch is configured to selectivelysupplement a reference current with current from the capacitor based onthe digital output signal to generate the analog feedback signal. 20.The device of claim 17, comprising the data storage location, whereinthe data storage location comprises a floating gate transistor, aresistive memory element, a photo-diode, or a combination thereof.